The present invention relates to a control apparatus for an elevator in which an induction motor for driving a cage is subjected to the variable voltage and variable frequency control (hereinbelow, termed "V.V.V.F. control"), and more particularly to improvements in the control performance of a vector control operation.
FIG. 1 is a block diagram showing the arrangement of a control apparatus of this type, along with a system for balancing a cage. Referring to the figure, a hoisting rope 3 is wound round a sheave 1 as well as a deflector wheel 2. A cage 4 is coupled to one end of this hoisting rope 3, and a balance weight 5 to the other end. Disposed below them is a tension pulley 6, a round which a compensation rope 7 is wound with its one end coupled to the cage 4 and its other end coupled to the weight 5. A brake 8 is disposed on the outer side of the sheave 1.
Here, the sheave 1 is coupled to an induction motor 9. In order to drive this induction motor 9, the control apparatus comprises a power conversion device which includes a thyristor converter 11 connected to a three-phase A.C. power source 10, a capacitor 12 for smoothing the rectified voltage of the thyristor converter, and a transistor inverter 13 for inverting the smoothed direct current into alternating current; and a voltage/frequency control device (hereinbelow, simply termed "control device") 20 which controls the power conversion device by receiving the current signal of a current sensor 14 for detecting the current of the induction motor 9 and the velocity signal of a velocity sensor (tachometer generator) 15 for sensing the rotational velocity of the induction motor 9.
In this case, the control device 20 has an interface (I/F), a read only memory (ROM), a random access memory (RAM) and a microprocessor (CPU). It receives and stores control command values besides the signals of the aforementioned sensors, and subjects the power conversion device to the pulse width modulation (PWM) control on the basis of the stored data. Thus, the A.C. voltage of an approximate sinusoidal wave having any desired voltage value and frequency value is applied to the induction motor 9.
With such control apparatus, when the ratio between the voltage and the fequency is maintained in a predetermined relationship, the output torque of the induction motor 9 can be kept constant. However, when it is intended to attain a control performance equivalent to that of a D.C. motor, the so-called vector control which controls also the phase is required.
FIG. 2 shows an example of arrangement of the control device which is called the "slip frequency type vector control system". Although the actual vector control is performed by the use of a microcomputer, the device is illustrated in model-like fashion here in order to facilitate understanding of the principle.
This control device 20a is composed of a velocity control amplifier 21, a differentiator 22, dividers 23a, 23b, coefficient units 24a-24e, a D.C. component vector computing unit 25, adders 26a, 26b, a vector oscillator 27, a vector multiplier 28, a vector three-phase converter 29, and operational amplifiers 30a, 30b, 30c.
Now, the outline of the slip frequency vector control will be described with reference to FIG. 2.
The slip frequency vector control evaluates a magnetic flux component current and a torque component current as values on secondary magnetic flux coordinates. When to be changed into primary current values on fixed coordinates, they are changed without detecting a secondary magnetic flux vector.
In this case, a slip frequency which is determined by the magnetic flux component current and the torque component current is calculated using motor constants. This slip frequency and a rotational velocity are subsequently added to find the rotational velocity of the secondary magnetic flux. This rotational velocity is further integrated to obtain a position, which is used for the coordinate transformation as the estimated position of the secondary magnetic flux.
Thus, the control device is supplied with a motor velocity signal as the state variable of the induction motor for the addition with the slip frequency, but it is not supplied with any signal concerning the secondary magnetic flux. The magnetic flux control does not perform the feedback control, either, and a first-order lead is directly calculated and obtained from a secondary magnetic flux command so as to compensate for the first-order lag response of the secondary magnetic flux to the magnetic flux component current.
Accordingly, the slip frequency vector control is a kind of predictive control which performs the control while estimating the magnetic flux vector by the calculations.
In FIG. 2, the control device 20a is supplied with the velocity signal .omega..sub.r of the velocity sensor 15 and a velocity command .omega..sub.r * as well as a secondary magnetic flux command .phi..sub.2 *. The velocity signal .omega..sub.r and the velocity command .omega..sub.r * are applied to the velocity control amplifier 21. A value obtained by amplifying the deviation between them is denoted as a torque command T.sub.M *. This torque command T.sub.M * is divided by a secondary magnetic flux command .phi..sub.2 * by means of divider 23a, to obtain a secondary q-axis current command -i.sub.2q *. This secondary q-axis current command -i.sub.2q * is multiplied by L.sub.2 /M by means of the coefficient unit 24a, to obtain a torque component current command i.sub.1q *. Here, L.sub.2 denotes a self-inductance of a secondary (rotor) winding, and M the mutual inductance between a primary (stator) winding and the secondary (rotor) winding.
After the secondary magnetic flux command .phi..sub.2 * is differentiated by the differentiator 22, it is multiplied by 1/R.sub.2 by means of the coefficient unit 24b. Further, the resulting product is multiplied by L.sub.2 /M by means of the coefficient unit 24c, and the product thus obtained is applied as an input to the adder 26a as current for forming the secondary magnetic flux proportional to a time variation rate. Here, R.sub.2 denotes the resistance of the secondary winding. On the other hand, the secondary magnetic flux command .phi..sub.2 * is multiplied by 1/M by means of the coefficient unit 24d, and the product thus obtained is applied as an input to the adder 26a as an exciting current for obtaining the secondary magnetic flux. The adder 26a sums both the inputs, and delivers as an output a magnetic flux component current command i.sub.1d * which compensates for the secondary magnetic flux with the first-order lag over the magnetic flux component current. A forming voltage must be generated across the secondary winding because of the indirect control of the secondary magnetic flux, and the above is intended to generate this voltage by causing a forming current to flow through the secondary winding.
Meanwhile, the secondary q-axis current command -i.sub.2q * is multiplied by R.sub.2 by means of the coefficient unit 24e and is subsequently divided by the secondary magnetic flux command value .phi..sub.2 * by means of the divider 23b, the resulting quotient being applied to the adder 26b as a slip frequency command .omega..sub.s *. This adder 26b adds the slip frequency command .omega..sub.s * and the aforementioned velocity signal .omega..sub.r, to evaluate the velocity command .omega..sub.0 * of the secondary magnetic flux and to apply it to the vector oscillator 27.
This vector oscillator 27 obtains a unit vector e.sup.j.theta.o* which indicates the predictive position .theta..sub.o * (=.intg..omega..sub.o * dt) of the secondary magnetic flux.
The magnetic flux component vector computing unit 25 obtains that primary current vector i.sub.1 *(.theta..sub.o *) on secondary magnetic flux coordinates which is determined by the torque component current command i.sub.1q * and the magnetic flux component current command i.sub.1d *.
The unit vector e.sup.j.theta.o* and the primary current vector i.sub.1 *(.theta..sub.o *) which have been thus obtained are multiplied by the vector multiplier 28, to be converted into a primary current vector i.sub.1 * on fixed coordinates, which is applied to the vector three-phase converter 29.
Here, the vector three-phase converter 29 determines the current command values i.sub.U *, i.sub.V * and i.sub.W * of the respective phases on the basis of the primary current vector i.sub.1 *.
These current command values are applied to the transistor inverter 13 through the operational amplifiers 30a, 30b and 30c, respectively.
Thus, the control device 20 shown in FIG. 1 performs the control substantially as stated above. In the slip frequency type vector control, the induction motor constants R.sub.2, L.sub.2 and M are necessary. Among them, the resistance R.sub.2 of the secondary winding has been handled as a constant value in spite of the fact that it changes conspicuously due to the temperature rise of the rotor.
Therefore, the prior-art control apparatus for the elevator has been less immune against disturbances and has been incapable of a high precision control.